僅需要少數廉價元器件，圖1中的電路斷路器響應過流和過壓故障。電路的核心處，一個可調高精度的并聯穩壓器D2，提供參考電壓、比較器和開集電極輸出，所有功能都整合到三管腳的封裝中。
圖2顯示ZR431, D1的簡化電路圖。在參考輸入處的電壓與內部電壓基準VREF相比較，名義上是2.5V。在斷電狀態下，參考電壓為0V，輸出晶體管處于截止狀態，陰極電流小于0.1µA。隨著參考電壓接近VREF，陰極電流緩慢增加；參考電壓超過2.5V的閾值時，裝置完全導通，陰極電壓降至大約為2V。在這種情況下，陰極和電源之間的阻抗決定陰極電流；陰極電流在50µA至100mA范圍內。
在正常工作情況下，D2的輸出晶體管截止，而且P溝道MOSFET（Q4）的門極通過R9，以至于MOSFET是全面增強的，允許負載電流ILOAD從電源–VS通過R6流到負載處。Q2和電流敏感電阻R6監測ILOAD的幅值，其中Q2的基極和發射極電壓VBE是ILOAD×R6。對于ILOAD的正常值，VBE低于Q2偏置所需的0.6V電壓值，
這種情況下晶體管對R3 和R4連結處的電壓也沒有影響。因為D2參考輸入的輸入電流小于1µA，通過R5可忽略壓降，而參考電壓實際上是R4上的電壓。
當ILOAD超過最大允許值時發生過載情況，R6上的電壓增大導致基極-發射極電壓足夠大到導通Q2。因此，R4上的電壓和參考電壓上拉到VS，造成D2的陰極電壓降至大約2V。D2的輸出晶體管通過R7 和 R8的瀉放電流，因此偏置Q3導通。Q4的柵極電壓通過Q3有效地控制電源，MOSFET從而截止。與此同時，Q3的源電流通過D1流到R4，從而拉動R4的電壓，使二極管電壓降到低于電源。由于Q2的基極-發射極的電壓為0V而截止，因此沒有負載電流流過R6。D2的輸出晶體管鎖存，電路仍處于故障狀態，其中的負載電流為0A。選擇R6的阻值時要確保在負載電流的最高允許值的條件下，Q2的基極-發射極電壓大約低于0.5V。
對于過流情況，該斷路器還對電源的非正常大電壓起作用。當負載電流在正常范圍內Q2處于截止狀態時，電源幅值以及R3和R4的值，穿過電源軌形成潛在分壓器，決定參考輸入處的電壓。電源電壓發生過壓情況，R4的電壓超過2.5V參考電壓，D2的輸出晶體管導通。一旦再次發生，Q3導通，MOSFET（Q4）關閉，負載與危險瞬間情況有效隔絕。
現在電路仍然處在不定狀態一直到復位。在這些條件下，Q3 控制 Q4的柵極電源電壓大約接近0V，從而保護MOSFET自身擺脫過多的柵源電壓。忽視R5微乎其微的電壓值，你可以看到參考電壓為VS×R4/(R3+R4)。因為，當參考電壓超過2.5V時，D2的輸出變高，你可以變換方程為R3=[(VST/2.5)–1]×R4，其中VST是所需的電源跳閘值。例如，如果R4值為10kΩ，18V的跳閘電壓需要R3阻值為62kΩ。R3 和 R4選擇適當的阻值來設置需要的跳閘電壓值，確保它們足夠大以至于潛在分壓器不會過度負荷供應。同樣，由于輸入參考電流避免導致誤差的值。
當你第一次對電路供電，會發現電容、燈泡燈絲、汽車等類似具有大浪涌電流的載荷可以使斷路器跳閘，即使正常的、穩態運行的電流低于R6所設置的水平。解決這個問題的一個辦法，就是增加電容C2，降低參考輸入處電壓的變化率。不過，雖然簡單，但這種方式有一個嚴重的缺點，因為它減緩了電路對于真正過流故障的響應時間。
器件C1,、R1,、R2和Q1提供了另一種解決方法。當電壓變大時，C1初始時放電，導致Q1導通，從而控制參考輸入為0V，防止來自跳閘電路的涌電流。然后C1通過R1 和R2 充電，直到Q1最后截止，釋放參考輸入的控制，并允許電路快速反應過流瞬變。此時C1、R1和 R2的值，電路允許涌電流在大約400毫秒內平息下來。選擇其它值可以使電路容納適用于負載的任何時限的涌電流。一旦你的電路斷路器跳閘，再次供電或者按下復位開關S1則可以復位。如果你的應用不需要涌電流保護，干脆省略C1、R1和 R2，并在參考輸入和0V之間接入S1。
在選擇元器件時，確保所有的元器件妥善適應它們將遇到的電壓和電流水平。雙極晶體管沒有特別的要求，雖然這些晶體管，尤其Q2和Q3，應具有高電流增益，Q4應該有較低的阻值，并且Q4的最大漏源極電壓和柵源極電壓必須與最高電源電壓相同。你可以為D1使用幾乎任何一個小信號的二極管。作為一項預防措施，如果有非常大的瞬態電壓，適當的齊納二極管D3 和D4保護D2可能是有必要的。
雖然該電路利用431器件，是市面從不同廠家都有的廣泛產品，對于D2，并不是所有這些產品都表現的一模一樣。舉例來說，測試了德州儀器的TL431CLP和Zetex公司的ZR431CLP，顯示當參考電壓為0V時兩個器件的陰極電流是0A。但是，逐步把參考電壓從2.2V提高到2.45V，對于TL431CLP而言，陰極電流由220提高到380µA，而ZR431CLP是從 23到28µA—兩者大概有10種區別。在選擇R7 和R8的阻值時，你必須考慮這兩種不同大小的陰極電流的區別。
你所使用的D2類型和你選擇的R7 和R8的阻值也對響應時間有影響。TL431CLP的一個測試電路，其中R7 是1kΩ，R8是4.7kΩ，對于瞬態過流的響應時間是550ns。用ZR431CLP更換TL431CLP，其響應時間約為1µs。增加R7 和R8的阻值分別到10和47kΩ，則響應時間為2.8µs。注意到TL431CLP產生較大的陰極電流需要相應阻值較小的R7和R8。
為了設定過壓跳閘水平在18V，R3 和 R4必須具備阻值分別為62和10kΩ。測試電路實驗得出如下結果：D2采用TL431CLP，電路在17.94V跳閘，D2采用ZR431CLP，跳閘電壓為18.01V。依靠Q2的基極-發射極電壓，過流檢測機制的精度低于過壓功能。然而，用一個高端電流檢測放大器產生與負載電流成正比的地電流來取代R6和 Q2，過流檢測精度將大大提高。這些器件可從Linear技術公司、Maxim、德州儀器公司和Zetex等公司得到。
電路斷路器被證明是很有用的應用，例如汽車系統，需要過流檢測，以防止錯誤載荷；還需要過壓保護，屏蔽敏感電路受到高能負載瞬變時的影響。除了流過R3 和R4的小電流，以及D2的陰極電流，在正常、非跳閘情況下，對于電源，電路沒有電流流出。
英文原文：
Circuit breaker provides overcurrent and precise overvoltage protection
A simple circuit breaker delivers precision overvoltage protection and overcurrent protection.
Anthony H Smith, Scitech, Bedfordshire, England; Edited by Brad Thompson and Fran Granville -- EDN, 6/7/2007
Requiring only a handful of inexpensive components, the circuit breaker in Figure 1 responds to both overcurrent- and overvoltage-fault conditions. At the heart of the circuit, D2, an adjustable, precision, shunt-voltage regulator, provides a voltage reference, comparator, and open-collector output, all integrated into a three-pin package.
Figure 2 shows a simplified view of the ZR431, D1. The voltage appearing at the reference input is compared with the internal voltage reference, VREF, nominally 2.5V. In the off state, when the reference voltage is 0V, the output transistor is off, and the cathode current is less than 0.1 µA. As the reference voltage approaches VREF, the cathode current increases slightly; when the reference voltage exceeds the 2.5V threshold, the device fully switches on, and the cathode voltage falls to approximately 2V. In this condition, the impedance between the cathode and the supply voltage determines the cathode current; the cathode current can range from 50 µA to 100 mA.
Under normal operating conditions, D2’s output transistor is off, and the gate of P-channel MOSFET Q4 goes through R9, such that the MOSFET is fully enhanced, allowing the load current, ILOAD, to flow from the supply voltage, –VS, through R6 into the load. Q2 and current-sense resistor R6 monitor the magnitude of ILOAD, where Q2’s base-emitter voltage, VBE, is ILOAD×R6. For normal values of
ILOAD, VBE is less than the 0.6V necessary to bias Q2 on, such that the transistor has no effect on the voltage at the junction of R3 and R4. Because the input current at D2’s reference input is less than 1 µA, negligible voltage drops across R5, and the reference voltage is effectively equal to the voltage on R4.
In the event of an overload when ILOAD exceeds its maximum permissible value, the increase in voltage across R6 results in sufficient base-emitter voltage to turn on Q2. The voltage on R4 and, hence, the reference voltage now pull up toward VS, causing D2’s cathode voltage to fall to approximately 2V. D2’s output transistor now sinks current through R7 and R8, thus biasing Q3 on. Q4’s gate voltage now effectively clamps to the supply voltage through Q3, and the MOSFET turns off. At the same instant, Q3 sources current into R4 through D1, thereby pulling the voltage on R4 to a diode drop below the supply voltage. Consequently, no load current flows through R6 because Q2, whose base-emitter voltage is now 0V, has turned off. As a result, no load current flows through R6, D2’s output transistor latches on, and the circuit remains in its tripped state in which the load current is 0A. When choosing a value for R6, ensure that Q2’s base-emitter voltage is less than approximately 0.5V at the maximum permissible value of the load current.
As well as responding to overcurrent conditions, the circuit breaker also reacts to an abnormally large value of the supply voltage. When the load current lies within its normal range and Q2 is off, the magnitude of the supply voltage and the values of R3 and R4, which form a potential divider across the supply rails, determine the voltage at the reference input. In the event of an overvoltage at the supply voltage, the voltage on R4 exceeds the 2.5V reference level, and D2’s output transistor turns on. Once again, Q3 turns on, MOSFET Q4 switches off, and the load becomes effectively isolated from the dangerous transient.
The circuit now remains in its tripped state until reset. Under these conditions, Q3 clamps Q4’s gate-source voltage to roughly 0V, thereby protecting the MOSFET itself from excessive gate-source voltages. Ignoring the negligibly small voltage across R5, you can see that the reference voltage is VS×R4/(R3+R4) in volts. Because D2’s output turns on when the reference voltage exceeds 2.5V, you can rearrange the equation as R3=[(VST/2.5)–1]×R4 in ohms, where VST is the required supply-voltage trip level. For example, if R4 has a value of 10 kΩ, a trip voltage of 18V would require R3 to have a value of 62 kΩ. When choosing values for R3 and R4 to set the desired trip voltage, ensure that they are large enough that the potential divider will not excessively load the supply. Similarly, avoid values that could result in errors due to the reference-input current.
When you first apply power to the circuit, you’ll find that capacitive, bulb-filament, motor, and similar loads having large inrush current can trip the circuit breaker, even though their normal, steady-state operating current is below the trip level that R6 sets. One way to eliminate this problem is to add capacitor C2, which slows the rate of change of the voltage at the reference input. However, although simple, this approach has a serious disadvantage in that it slows the circuit’s response time to a genuine overcurrent-fault condition. Components C1, R1, R2, and Q1 provide an alternative solution. On power- up, C1 initially discharges, causing Q1 to turn on, thereby clamping the reference input to 0V and preventing the inrush current from tripping the circuit. C1 then charges through R1 and R2 until Q1 eventually turns off, releasing the clamp at the reference input and allowing the circuit to respond rapidly to overcurrent transients. With the values of C1, R1, and R2, the circuit allows approximately 400 msec for the inrush current to subside. Selecting other values allows the circuit to accommo
date any duration of inrush current you apply to a load. Once you trip the circuit breaker, you can reset it either by cycling the power or by pressing S1, the reset switch, which connects across C1. If your application requires no inrush protection, simply omit C1, R1, R2, and Q1 and connect S1 between the reference input and 0V.
When choosing components, make sure that all parts are properly rated for the voltage and current levels they will encounter. The bipolar transistors have no special requirements, although these transistors, especially Q2 and Q3, should have high current gain, Q4 should have low on-resistance, and Q4’s maximum drain-to-source and gate-to-source voltages must be commensurate with the maximum value of supply voltage. You can use almost any small-signal diode for D1. As a precaution, it may be necessary to fit zener diodes D3 and D4 to protect D2 if extremely large transient voltages are likely.
Although this circuit uses the 431 device, which is widely available from different manufacturers, for D2, not all of these parts behave in exactly the same way. For example, tests on a Texas Instruments TL431CLP and a Zetex ZR431CL reveal that the cathode current is 0A for both devices when the reference voltage is 0V. However, gradually increasing the reference voltage from 2.2 to 2.45V produces a change in cathode current ranging from 220 to 380 µA for the TL431CLP and 23 to 28 µA for the ZR431CL—roughly a factor of 10 difference between the two devices. You must take this difference in the magnitude of the cathode current into account when selecting values for R7 and R8.
The type of device you use for D2 and the values you select for R7 and R8 can also have an effect on response time. A test circuit with a TL431CLP, in which R7 is 1 kΩ and R8 is 4.7 kΩ, responds within 550 nsec to an overcurrent transient. Replacing the TL431CLP with a ZR431CL results in a response time of approximately 1 µsec. Increasing R7 and R8 by an order of magnitude to 10 and 47 kΩ, respectively, produces a response time of 2.8 µsec. Note that the relatively large cathode current of the TL431CLP requires correspondingly small values of R7 and R8.
To set the overvoltage-trip level at 18V, R3 and R4 must have values of 62 and 10 kΩ, respectively. The test circuit then produces the following results: Using a TL431CLP for D2, the circuit trips at 17.94V, and, using a ZR431CL for D2, the trip level is 18.01V. Depending on Q2’s base-emitter voltage, the overcurrent-detection mechanism is less precise than the overvoltage function. However, the overcurrent-detection accuracy greatly improves by replacing R6 and Q2 with a high-side current-sense amplifier that generates a ground-referred current proportional to load current. These devices are available from Linear Technology, Maxim, Texas Instruments, Zetex, and others.
The circuit breaker should prove useful in applications such as automotive systems that require overcurrent detection to protect against faulty loads and that also need overvoltage protection to shield sensitive circuitry from high-energy-load-dump transients. Other than the small current flowing in R3 and R4 and the current in D2’s cathode, the circuit draws no current from the supply in its normal, untripped state.